1. Statement of the Technical Field
The inventive arrangements relate generally to methods and apparatus for providing increased design flexibility for RF circuits, and more particularly for localized optimization of the properties of dielectric circuit board materials for improved log-periodic dipole array (LPDA) antenna performance.
2. Description of the Related Art
RF circuits, transmission lines and antenna elements are commonly manufactured on specially designed substrate boards. Conventional circuit board substrates are generally formed by processes such as casting or spray coating which generally result in uniform substrate physical properties, including the dielectric constant.
For the purposes RF circuits, it is generally important to maintain careful control over impedance characteristics. If the impedance of different parts of the circuit do not match, signal reflections and inefficient power transfer can result. Electrical length of transmission lines and radiators in these circuits can also be a critical design factor.
Two critical factors affecting circuit performance relate to the dielectric constant (sometimes referred to as the relative permittivity or xcex5r) and the loss tangent (sometimes referred to as the dissipation factor) of the dielectric substrate material. The relative permittivity determines the speed of the signal in the substrate material, and therefore the electrical length of transmission lines and other components disposed on the substrate. The loss tangent characterizes the amount of loss that occurs for signals traversing the substrate material. Accordingly, low loss materials become even more important with increasing frequency, particularly when designing receiver front ends and low noise amplifier circuits.
Printed transmission lines, passive circuits and radiating elements used in RF circuits are typically formed in one of three ways. One configuration known as microstrip, places the signal line on a board surface and provides a second conductive layer, commonly referred to as a ground plane. A second type of configuration known as buried microstrip is similar except that the signal line is covered with a dielectric substrate material. In a third configuration known as stripline, the signal line is sandwiched between two electrically conductive (ground) planes.
Feed lines can also provide impedance transformations. For example, it is well known that a quarter-wavelength section of line can be designed to provide a match between a desired transmission line impedance and a given load impedance. For example, assuming the load and source impedances are substantially resistive, a transmission line can be matched to a load at the termination of the quarter-wave section if the characteristic impedance of the quarter wave section   Z      λ    4  
is selected using the equation:       Z          λ      4        =                    Z        01            ⁢              Z        02            
where   Z      λ    4  
is the characteristic impedance of the quarter-wave section;
Z01 is the characteristic impedance of the input transmission line; and
Z02 is the load impedance.
Simple quarter-wave transformers will operate most effectively only over a relatively narrow bandwidth where the length of the transformer approximates a quarter-wavelength at the frequency of interest. In order to provide matching over a broader range of frequencies, a multi-section transformer can be designed with a plurality of matching stages. For example, rather than attempting to use a single quarter-wave transmission line to transform from an impedance of 50 ohms to 10 ohms, one could use two quarter-wave sections in series. In that case, the first quarter wave section might be designed to transform from 50 ohms to 30 ohms, and the second quarter wave section might transform from 30 ohms to 10 ohms. Notably, the two quarter-wave sections when arranged in series would together comprise a half-wave section. However, this half wave section would advantageously function as a quarter-wave transformer section at half the design frequency. This technique can be used to achieve matching that is more broad-banded as compared to a simple quarter-wavelength section.
As the number of transformer stages is increased, the impedance change between sections becomes smaller. In fact, a transformer can be designed with essentially an infinite number of stages such that the result is a smooth, continuous variation in impedance Z(x) between feed line Z0 and load ZL. For maximally wide pass band response and a specified pass band ripple the taper profile can have an analytic form known as the Klopfenstein taper. There is substantial literature devoted to the design of multiple section and tapered transmission line transformers.
One problem with multiple transformer sections and tapered line transformers is that they are physically large structures. In fact, multiple section transformers are generally multi-quarter wavelengths long at the design frequency and tapered line transformers are generally at least about one wave-length long at the lowest design frequency and the minimum length is, to a degree, dependent on the impedance ratio. Accordingly, these designs are in many cases not compatible with the trend toward application of miniature semiconductors and integrated circuits.
Yet another problem with transmission line impedance transformers is the practical difficulties in implementation in microstrip or stripline constructions. For example, for a given dielectric substrate having a predetermined permittivity, the characteristic impedance of a transmission line is generally a function of the line width. Consequently, the width of the transformer section can become impractically narrow or wide depending on the transformation that a designer is trying to achieve, i.e., the impedance at each end of the transformer section.
In general, the characteristic impedance of a parallel plate transmission line, such as stripline or microstrip, is approximately equal to {square root over (L1/C1)}, where L1 is the inductance per unit length and C1 is the capacitance per unit length. The values of L1 and C1 are generally determined by the physical geometry and spacing of the line structure as well as the permittivity of the dielectric material(s) used to separate the transmission lines.
In conventional RF designs, a substrate material is selected that has a single relative permittivity value and a single relative permeability, the relative permeability value being about 1. Once the substrate material is selected, the line characteristic impedance value is generally exclusively set by controlling the geometry of the line.
The dielectric constant of the selected substrate material for a transmission line, passive RF device, or radiating element determines the physical wavelength of RF energy at a given frequency for that structure. One problem encountered when designing microelectronic RF circuitry is the selection of a dielectric board substrate material that is reasonably suitable for all of the various passive components, radiating elements and transmission line circuits to be formed on the board.
In particular, the geometry of certain circuit elements may be physically large or miniaturized due to the unique electrical or impedance characteristics required for such elements. For example, many circuit elements or tuned circuits may need to be an electrical xc2xc wave. Similarly, the line widths required for exceptionally high or low characteristic impedance values can, in many instances, be too narrow or too wide for practical implementation for a given substrate. Since the physical size of the microstrip or stripline is inversely related to the relative permittivity of the dielectric material, the dimensions of a transmission line can be affected greatly by the choice of substrate board material.
Still, an optimal board substrate material design choice for some components may be inconsistent with the optimal board substrate material for other components, such as antenna elements. Moreover, some design objectives for a circuit component may be inconsistent with one another. For example, it may be desirable to reduce the size of an antenna element. This could be accomplished by selecting a board substrate material with a high relative permittivity, such as 50 to 100. However, the use of a dielectric with a high relative permittivity will generally result in a significant reduction in the radiation efficiency of the antenna.
As with other components, an antenna design goal is frequently to effectively reduce the size of the antenna without too great a reduction in radiation efficiency. One method of reducing antena size is through capacitive loading, such as through use of a high dielectric constant substrate for the dipole array elements.
For example, if dipole arms are capacitively loaded by placing them on xe2x80x9chighxe2x80x9d dielectric constant board substrate portions, the dipole arms can be shortened relative to the arm lengths which would otherwise be needed using a lower dielectric constant substrate. This effect results because the electrical field in high dielectric substrate portion between the arm portion and the ground plane will be concentrated into a smaller dielectric substrate volume.
However, the radiation efficiency, being the frequency dependent ratio of the power radiated by the antenna to the total power supplied to the antenna, will be reduced primarily due to the shorter dipole arm length. A shorter arm length reduces the radiation resistance, which is approximately equal to the square of the arm length for a xe2x80x9cshortxe2x80x9d (less the xc2xd wavelength) dipole antenna as shown below:
Rr=20xcfx802(l/xcex)2
where l is the electrical length of the antenna line and xcex is the wavelength of interest.
A conductive trace comprising a single short dipole can be modeled as an open transmission line having series connected radiation resistance, an inductor, a capacitor and a resistive ground loss. The radiation efficiency of such a dipole antenna system, assuming a single mode, can be approximated by the following equation:   E  =            R      r              (                        R          r                +                  X          L                +                  X          C                +                  R          L                    )      
Where
E is the efficiency
Rr is the radiation resistance
XL is the inductive reactance
XC is the capacitive reactance
XL is the ohmic feed point ground losses and skin effect
The radiation resistance is a fictitious resistance that accounts for energy radiated by the antenna. The inductive reactance represents the inductance of the conductive dipole lines, while the capacitor is the capacitance between the conductors. The other series connected components simply turn RF energy into heat, which reduces the radiation efficiency of the dipole.
An inherent problem with the conventional substrate approach is that, at least with respect to the dielectric substrate, the only control variable for line impedance is selection of a single relative permittivity. This limitation highlights an important problem with conventional substrate materials, i.e. they fail to take advantage of the other factor that determines characteristic impedance, namely L1, the inductance per unit length of the transmission line. In addition, as noted above, conventional substrates do not provide the ability to vary the permittivity across the substrate area.
Yet another problem that is encountered in RF circuit design is the optimization of circuit components for operation on different RF frequency bands. Line impedances and lengths that are optimized for a first RF frequency band may provide inferior performance when used for other bands, either due to impedance variations and/or variations in electrical length. Such limitations can limit the effective operational frequency range for a given RF system.
Antenna elements are sometimes configured as antenna arrays, particular when broadband performance is desired. For example, a log-periodic dipole array (LPDA) represents a class of antennas in which a series of half-wavelength dipoles are arranged in a coplanar and parallel configuration on a transmission line. Such LPDAs are well known, and are in wide use. LPDAs are sometimes configured as an array of LPDAs and are commonly referred to as rose arrays.
The number of dipole elements used in an LPDA depends on the required performance characteristics. A metallic ground plane is generally located approximately one quarter-wavelength from each of the respective dipole elements.
An optimized LPDA would include a transmission line having feed line dimensions (length and width) that vary logarithmically along with the rest of the antenna dimensions, such as dipole length. Doing so, however, presents fabrication difficulties in realizing the required logarithmically varying dimensions. Thus, in practice, this form of the feed line is rarely seen because of fabrication difficulties.
Another shortcoming in conventional LPDAs also relates to the feed line. Feed lines are generally driven assuming they perform as microstrip lines having some impedance. To provide xc2xc wave electrical paths to ground for each dipole element, a non-planar structure is generally used, such as through use of a conically shaped ground plane. However, metal lines do not behave as microstrip lines as the distance from the feed line to the ground plane significantly increases. For example, excessive distances from ground can result as the feed line moves out from the feed point of an LPDA. Accordingly, conventional LPDA feed lines do not behave as a microstrip strip line beyond a small percentage (e.g. less than 30%) of the length of the feed line as measured from the feed point.
This non-ideal transmission line behavior can cause performance problems for the LPDA. The respective dipole elements of the LPDA are generally ideally spaced apart from one another such that a signal travelling along the transmission line flips about 180 degrees between dipole elements. However, since the feed line design can be substantially compromised, reasonable phasing of the respective elements in the LPDA may not be possible.
In addition, since the dipole elements are placed at roughly quarter wavelength over the ground in order to maintain some semblance of a constant impedance across the frequency range of the circuit (e.g. across roughly three octaves), the radiation pattern of each LPDA in a rose array is directed to the side and away from the axis of the array. Therefore, the resulting summed pattern from the LPDAs comprising the rose array is not optimized.
Accordingly, the use of conventional substrate boards which provide a single uniform dielectric material result in performance degradation for RF circuits in general, with LPDA-based circuits suffering additional performance degrading effects. Attempts to reduce the size of such circuits generally result in further degradation of circuit performance.
A printed circuit antenna array includes a plurality of log periodic dipole arrays (LPDAs). Each LPDA includes dipole elements with arms having reduced size through use of high effective permittivity substrate portions. The radiation efficiency degradation generally associated with use of a high permittivitty substrate can be be reduced through addition of magnetic particles to provide enhanced permeability in the high permittivity regions. The substrate preferably includes meta-materials.
The array includes a dielectric circuit board substrate, the substrate having at least a first portion, the first portion providing at least one of a first relative permeability and a first relative permittivity. The first relative permeability and first relative permittivity are different from a bulk portion of the substrate. The LPDAs are disposed on the substrate, each LPDA including at least one feed line and a plurality of dipole elements electrically connected to the feed line, wherein at least a portion of the dipole elements are disposed on the first portion.
The first relative permittivity can be at least 10. The first relative permeability can be at least 2, or from about 4 to 116. The first relative permeability is selected for increasing the radiation efficiency of the LPDAs as compared to the radiation efficiency resulting from use of a first permeability of about 1. The first relative permeability is preferably approximately equal to the square root of the first relative permittivity.
At least a portion of the feed lines are disposed on a second portion of the substrate, the second portion providing at least one of a second relative permeability and a second relative permittivity which are different from the bulk substrate. The feed lines can have electrical width that increases substantially logarithmically outward from at least one feedpoint of the LPDAs, even where the physical width of the feed lines are not substantially logarithmic, such as linear.
The second second relative permittivity can be at least 10. The second relative permeability can be at least 2, or from about 4 to 116. The second relative permittivity and permeability can be different as compared to the first relative permittivity and permeability.
The feed lines can function as a broadband impedance transformer. The broadband tranformer can include a plurality of segments. The plurality of segments can provide quarter wave electrical lengths, the respective electrical lengths determined at the highest frequency over which respective impedance transforms are to occur.
At least one of the second relative permittivity and second relative permeability can vary along a length of the feed lines. In this embodiment, the characteristic impedance of the feed lines can vary along their length in accordance with a tapered line type transformer. For example, the characteristic impedance of the feed lines can be at least partially determined by a gradation of at least one the second relative permittivity and second relative permeability along a length of the feed lines. The gradadation can continuously vary along at least a portion of the length of the feed lines.
The array can be substantially planar formed from a substrate having a substantially uniform thickness and including a substantially planar ground plane disposed beneath the substrate. The relative permittivity of the substrate beneath the feed lines can increase the feed lines extended from respective feed points. Thus, although the physical distance from the respective dipole elements to the ground plane is essentially the same for each dipole, the electrical distance is different. The relative permittivity increase can be linearly graded or can increase in steps.